Delta-sigma modulators with integral digital low-pass filtering

ABSTRACT

A delta-sigma modulator of an order of at least three having a signal transfer function defining a signal low-pass response, the signal transfer function has a numerator and a denominator, the denominator defined by a product of first and second functions defining respective first and second pole sets such that a roll-off frequency of a low-pass response defined by the first pole set is at least twice a roll-off frequency of a low-pass response defined by the second pole set. The delta-sigma modulator also has a noise transfer function defining a noise high-pass response, the noise transfer function having a numerator and a denominator, the denominator defined by the product of the first and second functions. The numerator of the noise transfer function differs from the numerator of the signal transfer function and characterizes the noise high-pass response.

CROSS REFERENCE TO RELATED APPLICATIONS

This application for patent is a continuation of the following application for patent:

Currently pending U.S. application Ser. No. 10/151,322, filed May 20, 2002 by Melanson and Yi, for Delta-Sigma Modulators with Integral Digital Low-Pass Filtering.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates in general to delta sigma modulators and in particular to delta sigma modulators with integral low-pass filtering.

2. Description of the Related Art

Delta-sigma modulators are particularly useful in digital to analog and analog to digital converters (DACs and ADCs). Using oversampling, the delta-sigma modulator spreads the quantization noise power across the oversampling frequency band, which is typically much greater than the input signal bandwidth. Additionally, the delta sigma modulator performs noise shaping by acting as a lowpass filter to the input signal and a highpass filter to the noise; most of the quantization noise power is thereby shifted out of the signal band.

The typical delta sigma modulator-based DAC includes a summer summing the input signal with negative feedback, a linear filter, a quantizer which driving the feedback loop, and a DAC generating the analog signal from the quantizer output. In a first order modulator, the linear filter comprises a single integrator stage while the linear filter in a higher order modulator comprises a cascade of a corresponding number of integrator stages. The quantizer can be either a one-bit or a multiple-bit quantizer. Higher-order modulators have improved quantization noise transfer characteristics over those of lower order modulators, although stability becomes a critical factor as the order increases.

Some applications, such as Super Audio Compact Disc (SACD) one-bit audio, are subject to significant out-of-band noise and therefore require either an external analog filter at the modulator output or an internal digital decimation filter. The analog filtering approach is expensive to implement with high accuracy and adds complexity to the overall design. Among other things, expensive precision components are required to obtain the Q factor defining the required flatness of the filter response in the proximity of the corner frequency. Digital filtering eliminates the problems of designing and building analog filters, but at the expense of significantly increased computational intensity.

Hence, some other technique is required if cost and simplicity are target design goals for delta sigma modulator data converters for use in applications with significant out-of-band noise.

SUMMARY OF THE INVENTION

The principles of the present invention are embodied in noise shapers, including delta-sigma modulators and delta-sigma data converters having integral digital low pass filtering. According to one embodiment, a feedback noise shaper is disclosed having an order of at least three. A first pole set defines a signal transfer function of a selected corner frequency and a second pole set having at least one pole at a frequency of at least twice the selected corner frequency defines a noise transfer function.

According to the inventive concepts, the signal transfer function of the delta-sigma modulator is carefully selected to provide low pass filtering of an input data stream. Consequently, high-Q analog filtering and/or computationally intensive digital filtering are no longer required. This delta-sigma modulator is particularly useful in applications such as one-bit audio which are subject to significant out-of-band noise and normally require external analog or internal digital filtering. The inventive concepts are demonstrated with respect to a sixth order system, built from a fifth-order Butterworth filter and a first-order higher frequency filter in a distributive feedback delta-sigma modulator, although other filter responses and/or modulator architectures may be used.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which:

FIG. 1 is a high level functional block diagram of a Super-Audio Compact Disc audio system demonstrating a use of the analog to digital converter of FIG. 2;

FIG. 2 is a functional block diagram of an exemplary delta-sigma data converter suitable for use in the analog to digital converter of FIG. 1A;

FIGS. 3A–3D are plots of the signal and noise transfer functions for the delta-sigma data converter of FIG. 2 for selected sets of coefficients and selected orders;

FIGS. 4A and 4B are plots in the z-plane of the poles and zeros of the noise transfer function for the preferred and an alternate embodiment delta-sigma data converter, respectively; and

FIG. 5 is a flow chart illustrating a procedure for designing a noise-shaper with integral signal filtering for a delta-sigma data converter according to the inventive concepts.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The principles of the present invention and their advantages are best understood by referring to the illustrated embodiment depicted in FIGS. 1–3 of the drawings, in which like numbers designate like parts.

FIG. 1 is a diagram of a typical system application of 1-bit digital to analog converter (DAC) 100 according to the principles of the present invention. In this example, DAC subsystem 100 forms part of an audio component 101, such as a compact disk (CD) player, digital audio tape (DAT) player or digital video disk (DVD) unit. A digital media drive 102 recovers the digital data, for example 1-bit audio data in the Sony/Philips 1-bit format from the given digital data storage media, and passes those data, along with clocks and control signals, to DAC subsystem 100. The resulting analog (audio) data undergoes further processing in circuit block 103 prior amplification in amplifier block 104. Amplifier block 104 then drives a set of conventional speakers 105, a headset or the like.

Multi-bit digital audio data is received word-serial through the SDATA pin timed by the sampling clock (SCLK). The left and right channel data are alternately processed in response to the left-right clock (LRCK). This clock is normally at the same rate as the data input rate (i.e., the sampling rate). Control signals DF1 and DF0 allow for the selection of the input format, for example, right or left justified, 20-bit or 24 bit word width, to name a few possibilities. When 1-bit data is being input, the SDATA port receives left channel data and the DF1 port channel data.

FIG. 2 is an exemplary 6th order distributed-feedback delta-sigma modulator 200 based on six (6) integrator stages 201 a–201 f and associated summers 202 a–202 f and two local resonator feedback loops. The loop-filter equations and corresponding NTF and STF for this type of modulator, as well as for alternate configurations suitable for practicing the inventive principles, can be derived from the discussions of Norsworthy et al, Delta-Sigma Converters, Theory, Design and Simulation, IEEE Press (1997).

Quantizer 203 quantizes the modulated signal from the final integrator stage 201 f. Quantizer 203 is typically a multiple-bit quantizer. The quantized data in turn drive a high-speed, low-resolution output DAC 204, for example a sixteen (16)-level DAC, which generates the analog output signal. The input data, such as one-bit audio or PCM data out of an interpolator, can be input directly into first stage 201 a of the modulator 200 or through an optional finite impulse response (FIR) filter 205 described further below.

Typical delta sigma modulators are designed with a noise transfer function (NTF) for optimizing the noise shaping response described above. The associated signal transfer function (STF) is set to approximately unity gain or equalized or worked around elsewhere in the circuit. Normally, sufficient out-of-band gain is selected to drive in-band noise down to an acceptable level, typically 10 dB below the analog noise. The present inventive concepts however take advantage of the STF for filtering out-of-band noise in the input signal.

For discussion purposes, that one-bit audio data is assumed being processed at a 128*44100 Hz (5.6448 MHz) oversampling rate. Based on this assumption, the following exemplary filter coefficients, noise, signal transfer functions were developed. The data rate, the filter coefficients, and the transfer functions may vary from embodiment to embodiment without deviating from the inventive concepts. Moreover, the order and/or the configuration of the delta sigma modulator may vary from the preferred embodiment shown in FIG. 2. (For example, a feedforward or other known delta-sigma modulator architectures may be used in the alternative).

If a 50 kHz filter were used for the STF, the NTF would have insufficient in-band noise attenuation. Therefore, in modulator 200, the coefficients are selected to implement a STF filter with 50 kHz 5^(th) order Butterworth low pass poles and an additional pole at approximately 600 kHz. (See FIGS. 4A and 4B). (The characteristics of Butterworth low pass filters are described in most texts on digital signal processing, for example Proakis et al., Digital Signal Processing Principles, Theory and Applications, Prentice Hall, (1996)). The 600 kHz pole ensures that the NTF noise attenuation in the low pass band is sufficiently high. The coefficients for this embodiment are shown in

TABLE 1 TABLE 1 FEEDBACK COEFFICIENTS 5^(th) Order Butterworth Poles, 50 kHz corner, 128 fs, fs = 44.1 Coefficient kHz, Additional Pole at 600 kHz C1 2.51692 × 10⁻⁷ C2 0.0000152454 C3 0.00042963 C4 0.00811408 C5 0.0994375 C6 0.595056

One method of deriving these coefficients is through the Mathmatica® code provided in the appendix. FIGS. 3A–3D are exemplary plots for the NTF and STF generated by this code for the coefficients and constraints of Table 1. Specifically, the signal transfer function is shown in FIGS. 3A and 3B, and FIG. 3B zooming-in on the low frequency signal pass band. As shown, the passband corner is at approximately 50 kHz and the gain rolls-off relatively quickly thereafter. Similar plots for the NTF are shown in FIGS. 3C and 3D. With respects to the NTF, good noise attenuation is provided in the audio band up to approximately 20 kHz. The modulator structure of FIG. 2, which includes two local resonator feedback loops S1 and S2, results in two zeros in the audio passband.

Referring back to FIG. 2, a non-volatile or volatile memory 206, or equivalent logic, may be included to support programmability of the modulator coefficients, and hence selectability of the STF pass-band. For example, one set of low-frequency coefficients could be stored for the one-bit audio application, and another set of high-frequency coefficients for 192 kHz PCM applications with a corner frequency around 200 KHz. Multiplexers 208 a–208 f or similar selector-switching circuitry are utilized for performing the coefficient selection. Additionally, since a change in coefficients may change the scale factors for each stage, a pair of corresponding amplifiers 209 a–209 f and 210 a–210 f and associated multiplexers 211 a–211 f are provided in front of each stage.

Optional FIR 205 may be a simple design with five zeros at the Nyquist frequency. If any additional output filtering is required, a simple analog R-C filter at the data converter output may be utilized, In any event, by embodying the primary filter function in the STF of the delta sigma modulator, the overall system is much less expensive and/or much less computationally intensive.

Additionally, in audio applications, volume control can be implemented with a relatively simple one-bit by n-bit multiplier 207 operating on the one-bit audio data at the front end of the converter.

A plot of the zeros and poles in the z-plane for the NTF in the preferred embodiment is provided in FIG. 4A. The poles of the STF are co-located with those of the NTF and hence have not been separately plotted. According to the inventive principles, two distinct sets of poles are used. For the exemplary sixth (6^(th)) order Butterworth filter and the coefficients provided in Table 1, the first pole set, providing the 50 kHz corner frequency and a relatively sharp roll-off, includes one (1) real pole and two complex-conjugate pole pairs at z=0.945838, 0.981579±0.0520094i, and 0.955454±0.0312881i. The additional pole (second pole set) for increasing the noise attenuation in the signal passband is set at z=0.48497 or approximately 600 kHz. The zeros of the NTF are generally shown on the unit circle with two of six zeros moved off the real axis by the resonator loops of FIG. 2.

Generally, the first set of poles is selected to set the STF corner frequency (here approximately 50 kHz) and the slope (roll-off) at the passband edge of the low-pass signal filter. While one (1) real and two complex pole-pairs are used to set the STF characteristics in this example, the number and type of poles can change from embodiment to embodiment. Preferrably, the first pole set has at least two poles to define the STF which are preferrably complex to obtain a sufficiently sharp passband roll-off. While these poles are at the Butterworth locations, other pole locations (filter functions) and filter types, such as those used to obtain elliptical and Chebyshev functions can be used in the alternative.

The second pole or second set of poles is selected to improve the noise shaping function of the NTF, and particularly to increase the noise attenuation within the STF passband, as described above. Advantageously, by moving the second set of poles relative to the first set, the NTF can be tuned. In the illustrated embodiment, the second pole set comprises one real pole corresponding to a frequency of at least twice the STF cut-off frequency set by the first pole set (here 600 kHz). By moving this pole outward from the origin in the z-plane by reformulating the coefficient set, and therefore reducing the spacing of the first and second pole sets, the noise attenuation in the low frequency band will be reduced. On the other hand, the low frequency band NTF can generally be improved by moving this pole set towards the origin. The illustrated single real pole can moved to approximately the origin by when the coefficient of the last loop filter stage is approximately one (1).

As previously indicated, numerous alternate arrangements of the poles, as well as the zeros are possible, as illustrated in FIG. 4B. In this case, a sixth (6^(th)) order modulator is represented with one real pole and one complex pole-pair set the SFT low-response and one real pole and one complex pole-pair set the NTF.

FIG. 5 is a flow chart 500 of a procedure, which along with the code provided in the Appendix, provide a methodology for designing a delta-sigma data converter with integral signal filtering according to the inventive concepts.

At Blocks 501 and 502, show steps of a modulator structure being selected and the equations for the corresponding NTF and STF being derived. Exemplary structures and characterizing equations are set out in Norsworthy et al., cited in full above. Again, in the illustrated embodiment, a feedback structure was selected with two resonators; equations for the NTF and STF for this structure are provided in the code set out in the Appendix.

Block 503 shows the step of the desired low frequency response being selected for some initial number of loop-filter stages. From this selection, the initial desired transfer function is selected at block 504 and a determination is made as to whether this transfer function meets the in-band noise attenuation requirements for the signal at block 505. If it does not meet this requirement, then additional loop filter stages are added at block 506.

When the desired transfer function is achieved, the STF is defined to be equivalent to that of the transfer function at block 507. The coefficients are thereafter obtained at Block 508 by solving the equations for the NTF and STF for the selected structure and with the STF defined as the desired transfer function.

Although the present invention and its advantages have been described in detail, it should be understood that various changes, substitutions and alterations can be made herein without departing from the spirit and scope of the invention as defined by the appended claims.

APPENDIX In[20]:= sdanalyze [ord_, ds_, nz_] := Module [{c0, gg, a, bx, bn, cx, cn, dx, dn, izma, ceq, invm, hnn, hnx, afb, bfb}, (* Analyze a delta sigma topology ord - order ds - ds description function nz - number of complex zero pairs Returns characteristic equation (poles), NTF numerator, STF numerator, state transition matrix for feedback, and b vector for feedback *) co = Table [c[i], {i, ord}]; gg = Table [g[i], {i, nz}]; a = Transpose [Table[ Module [{i=Table[1f[j == k, 1, 0], {k, ord}]}, ds[i, 0, 0, co, gg, 0][[1]]] , {j, ord}]]; bx = Module [{i= Table [0, {ord}]}, ds[i, 1, 0, co, gg, 0] [[1]]]; bn = Module [{i = Table [0, {ord}]}, ds[i, 0, 1, co, gg, 0] [[1]]]; cx = cn = Table [Module[{i = Table [If[j== k, 1, 0], {k, ord}]}, ds [i, 0, 0, co, gg, 0] [[2]]] , {j, ord}]; dx = Module [{i = Table [0, {ord}]}, ds[i, 1, 0, co, gg, 0] [[2]]]; dn = Module [{i = Table [0, {ord}]}, ds[i, 0, 1, co, gg, 0] [[2]]]; izma = Table [If[j == k, 1, 0], {j, ord}, {k, ord}] − zm*a; ceq = Collect [Det [izma], zm]; hnn = Simplify [dn*ceq+zm* (cn.Table[ Simplify [Det[Transpose[ReplacePart[Transpose[izma], bn, i]]]], {i, ord]])] hxn = Simplify [dx*ceq+zm* (cx.Table[Simplify[ Det [Transpose[ReplacePart[Transpose[izma], bx, i]]]], {i, ord}])]; izma = .; afb = Transpose[Table[ Module [{i = Table[If[j == k, 1, 0], {k, ord}]}, ds[i, 0, 0, co, gg, 1] [[1]]] , {j, ord}]]; bfb = Module [{i = Table[0, {ord}]}, ds [i, 0, −1, co, gg, 0] [[1]]]; {ceq, hnn, hxn, afb, bfb} ]; In[21]:= solvecoef [fs_, fc_, n_, ceq_, zeros_, sp_] := Module [{1p, pp, ceqc, bnc}, (* Given a denominator polynomial, solve for the coeficients that give a butterworth responsed. sample freq, corner freq, order, poly in zm, local resonator feedbacks *) Ip[fsl_, fc1_, n1_] := Module [{cp, cp1, pz, r, a1, r1}, (* Make the z transform for a butterworth filter fsl - sample rate fcl - corner frequency n1 - order *) cp1 = Together[(1−zm* (1+s1)/(1 − s1))*(1 − zm * (1 + s2)/(1 − s2)) /. {s1 −> r1*Cos [x] + r*r1*Sin[x], s2 −> r1*Cos [x] − I*r1*Sin[x]}]; cp = Expand [Numerator[cp1]]/Expand[Denominator[cp1]]; rp = 1 − 1. * zm + (1 + s1)/(1 − s1)/.s1 −> − r1; (*.9999 to flatten freq response*) r := N[Tan [fc1/fs1*2*π/2]]; pz := Product [cp/.x−>iπ/(2 n1), {i, n1+1, 2 n1−1, 2}]/; EvenQ[n1]; pz := Product [cp/.x−>iπ/n1, {i, (n1+1)/2, n1−1}]*rp/; OddQ[n1]; N[Collect[pz/.{r1−> r}, zm]] ]; pp = If[sp == 1, 1p[fs, fc, n], 1p[fs, fc*sp, 1]*1p[fs, fc, n−1]]; ceqc = Flatten[Table[Coefficient[ ceq/.Table[g[i] −> zeros[[i]], {i, Length[zeros]}], zm{circumflex over ( )}i], {i, n}]]; bnc = Table[Coefficient[pp, zm{circumflex over ( )}i], {i, n}]; First[ Table[c[i], {i, n}]/.Solve[Inner[Equal, ceqc, bnc, List], Table[c[i], {i, n}]]] ]; In[22]: = makeh[den_, num_, coef_, ze_] := Module[{s1, s2}, s1 = Table[g[i] −> ze[[i]], {i, Length[ze]}); s2 = Table[c[i] −> coef[[i]], {i, Length[coef]}]; Simplify[num/.s1/.s2]/Simplify[den/.s1/.s2] ] In[23]: = patrsp[fs_, ntf_, stf_] := Module[{res, ww, sres}, res[x_] = ntf/.zm−>x; sres[x_] = stf/.zm−>x; Print[“Noise Transfer Function=\n”, ntf]; Print[“Signal Transfer Function=\n”, stf]; ww = N[2*π*I/fs]; Print[“Sample Rate= ”, fs]; Print[”Nyquest noise gain = ”, res [−1]]; Print[“noise reduction to 20k = ”, 10*Log[10,NIntegrate[Abs[res[Exp[ww*x]]]{circumflex over ( )}2, {x, 20, 20000}]/(fs/2)]]; Plot[20*Log[10, Abs[res[Exp[ww*x]]]], {x, 0, fs/2}, PlotRange −> {−120, 20}, GridLines −> Automatic, ImageSize −> 72*5, PlotLabel → “NTF”]; Plot[20*Log[10, Abs[sres[Exp[ww*x]]]], {x, 0, fs/2}, PlotRange −> {−120, 20}, GridLines −> Automatic, ImageSize −> 72*5, PlotLabel → “STF”]; Plot[20*Log[10, Abs[res[Exp[ww*x]]]], {x, 0, fs/32}, PlotRange −> {−120, 20}, GridLines −> Automatic, ImageSize −> 72*5, PlotLabel → “NTF ZOOM”]; Plot [20*Log[10, Abs[sres[Exp[ww*x]]]], {x, 0, fs/32}, PlotRange −> {− 120, 20}, GridLines −> Automatic, ImageSize −> 72*5, PlotLabel → “STF ZOOM”]; ]; In[24]: 5^(th) Order DSD re-modulator (*Define the topology of a modulator ii = starting state (list) x = input (scalar) ns = quantization “noise” (scalar) co = list of coeffecients g = local resonator gains fb = 1 to close loop, 0 open *) ds5[ii_, x_, ns_, co_, g_, fb_] := Module [{i0, i1, i2, i3, i4, y}, i0 = ii[[1]]; i1 = ii[[2]]; i2 = ii[[3]]; i3 = ii[[4]]; i4 = ii[5]]; y = If[fb == 1, ns, i4=ns]; i0 += (x−y) * co[[1]]; i1 += i0 + i2*g[[1]] − y*co[[2]]; i2 += i1 − y+co[[3]]; i3 += i2+i4*g[[2]] − y*co[[4]]; i4 += i3 − y*co[[5]]; {{i0, i1, i2, i3, i4}, y}   ]; In[25] := desc5 = sdanalyze[5, ds5, 2]; g1 = {−1./8192, −1./2048}; (*NTF zeros*) fs = 44100*128; ctab5 = solvecoef[fs, 50000, 5, desc5[[1]], g1, 1]; ntf5 = makeh[desc5[[1]], desc5[[2]], ctab5, g1]; stf5 = makeh[desc5[[1]], desc5[[3]], ctab5, g1] * ((.5+.5*zm){circumflex over ( )}1); In[31] := 6^(th) Order DSD re-modulator ds6[ii_, x_, ns_, co_, g_, fb_] := Module [{i0, i1, i2, i3, i4, i5, y}, i0 = ii[[1]]; i1 = ii[[2]]; i2 = ii[[3]]; i3 = ii[[4]]; i4 = ii[[5]]; i5 = ii[[6]]; y = If[fb == 1, ns, i5 + ns]; i4 += i3 + i5*g[[3]] − y*co[[5]]; i5 += i4 − y*co[[6]]; i2 += i1 + i3*g[[2]] − y*co[[3]]; i3 += i2 − y*co[[4]]; i1 += i0 + (−y) * co[[2]]; i0 += (x−y) co[[1]] + i1*g[[1]]; {{i0, i1, i2, i3, i4, i5}, y}   ]; In[32] := desc6 = sdanalyze[6, ds6, 3]; gl = {0, −2, {circumflex over ( )}−12, −2.{circumflex over ( )}−11); (*NTF zeros*) ctab6 = solvecoef[fs, 50000, 6, desc6[[1]], g1, 12]; ntf6 = makeh[desc6[[1]], desc6[[2]], ctab6, g1]; stf6 = makeh[desc6[[1]], desc6[[3]], ctab6, g1]; In[37] := Print [“coeficients = ”, ctab5]; patrsp[fs, ntf5, stf5]; Print [“coeficients = ”, ctab6]; patrsp[fs, ntf6, stf6]; coeficients = {4.88599×10⁻⁷, 0.0000259205, 0.000779239, 0.0138403, 0.164839} ${{Noise}\mspace{14mu}{Transfer}\mspace{14mu}{Function}} = \frac{\begin{matrix} {1.19737\left( {{- 1} + {zm}} \right)\left( {1 - {1.99988\mspace{20mu}{zm}} + {zm}^{2}} \right)} \\ \left( {1 - {1.99951\mspace{14mu}{zm}} + {zm}^{2}} \right. \end{matrix}}{\begin{matrix} {\left( {{- 1.05726} + {zm}} \right)\left( {1.09425 - {2.091\mspace{14mu}{zm}} + {zm}^{2}} \right)} \\ \left( {1.03498 - {2.03183\mspace{14mu}{zm}} + {zm}^{2}} \right) \end{matrix}}$ ${{Signal}\mspace{14mu}{Transfer}\mspace{14mu}{Function}} = \frac{5.85035\; \times 10^{- 7}\left( {0.5 + {0.5\mspace{14mu}{zm}}} \right){zm}}{\begin{matrix} {\left( {{- 1.05725} + {zm}} \right)\left( {1.09425 - {2.091\mspace{14mu}{zm}} + {zm}^{2}} \right)} \\ \left( {1.03498 - {2.03183\mspace{14mu}{zm}} + {zm}^{2}} \right) \end{matrix}}$ Sample Rate = 5644800 Nyquest noise gain = 1.09408 noise reduction to 20k = −85.2997 

1. A delta-sigma modulator comprising: a delta-sigma modulator of an order of at least three and having operating characteristics defined by: a signal transfer function defining a signal low-pass response, the signal transfer function having a numerator and a denominator, the denominator defined by a product of first and second functions defining respective first and second pole sets such that a roll-off frequency of a low-pass response defined by the first pole set is at least twice a roll-off frequency of a low-pass response defined by the second pole set; and a noise transfer function defining a noise high-pass response, the noise transfer function having a numerator differing from the numerator of the signal transfer function and defining characteristics of the noise high-pass response and a denominator, the noise transfer function denominator defined by the product of the first and second functions.
 2. The delta-sigma modulator of claim 1, wherein the numerator of the signal transfer function defines a signal transfer function zero.
 3. The delta-sigma modulator of claim 1, wherein the numerator of the signal transfer function defines a pure delay.
 4. The delta-sigma modulator of claim 1, wherein the numerator of the noise transfer function defines a noise transfer function zero.
 5. The delta-sigma modulator of claim 1, wherein the second function of the denominator of the signal transfer function defines the second pole set to produce a Butterworth lowpass filter response.
 6. The delta-sigma modulator of claim 5, wherein the second pole set has an order of five.
 7. The delta-sigma modulator of claim 1, wherein zeros of the noise transfer function are selected to define the noise high pass response.
 8. A method of integrating signal filtering and noise shaping into a delta-sigma modulator comprising: defining an order of the delta-sigma modulator to be at least three; defining a signal transfer function setting a modulator low-pass filter response for filtering an input signal having a selected baseband comprising: defining a numerator of the signal transfer function; and defining a denominator of the signal transfer function by a product of first and second functions, the first and second functions defining respective first and second pole sets such that a roll-off frequency of a low-pass response defined by the first pole set is at least twice a roll-off frequency of a low-pass response defined by the second pole set; and defining a noise transfer function setting a modulator noise high-pass response for removing noise from the selected baseband comprising: defining a numerator of the noise transfer function, the numerator of the noise transfer function differing from the numerator of the signal transfer function and defining characteristics of the modulator noise high-pass response; and defining a denominator of the noise transfer function by the product of the first and second functions.
 9. The method of claim 8, wherein defining the numerator of the signal transfer function comprises defining the numerator of the signal transfer function to introduce a pure delay into the signal transfer function.
 10. The method of claim 8, wherein defining the numerator of the signal transfer function comprises defining the numerator of the signal transfer function to introduce a zero into the signal transfer function.
 11. The method of claim 8, wherein defining the denominator of the signal transfer function comprises defining the first function to produce the second pole set producing a Butterworth lowpass response.
 12. The method of claim 11, wherein the second function produces a fifth order second pole set.
 13. The method of claim 8, wherein defining the numerator of the noise transfer function comprises defining the numerator of the noise transfer function to introduce a zero into the noise transfer function.
 14. The method of claim 8, wherein defining the numerators of the noise and signal transfer functions comprise defining numerators of the noise and signal transfer functions to introduce zeros into the noise and signal transfer functions. 